Motor and drive control device therefor

ABSTRACT

The invention provides a drive control device for the motor as well as an electric power steering apparatus using the motor and the drive control device. Respective phase current command values are calculated on the basis of vector control. Pseudo-vector control for controlling respective phases separately is used as current feedback control.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a divisional of and claims the benefit of priorityunder 35 U.S.C. §120 from the U.S. Ser. No. 10/536,990, filed May 31,2005, and claims the benefit of priority under 35 U.S.C. §119 fromJapanese Patent Application Nos. 2002-345135, 2002-354632 and2003-376428 filed Nov. 28, 2002, Dec. 6, 2002 and Nov. 6, 2003respectively, the entire contents of each which are incorporated hereinby reference.

TECHNICAL FIELD

The present invention relates to improvement of a motor, which is mostsuitable for use in an electric power steering apparatus, and a drivecontrol device for the motor as well as an electric power steeringapparatus using the motor and the drive control device.

BACKGROUND ART

Conventionally, a motor used in an electric power steering apparatus is,in general, a permanent magnet synchronous motor (PMSM), which is drivenby a three-phase sinusoidal current. As a control system for driving themotor, a control system called vector control is widely used. However,since there is a strong demand for a reduction in size of the electricpower steering apparatus, a brushless DC motor tends to be used as amotor suitable for the reduction in size.

Under such circumstances, a motor drive control device using the advanceangle control system for the conventional motor for the electric powersteering apparatus will be explained with reference to FIG. 1.

In a structure of the motor drive control device, a main path leading tothe motor 1 is connected to the back of a current command valuecalculating unit 100 that controls an electric current of a motor 1 viasubtracters 20-1, 20-2, and 20-3 that detect errors between phasecurrent command values Iavref, Ibvref, and Icvref and motor currents Ia,Ib, and Ic, a PI control unit 21 that inputs respective error signalsfrom the subtracters 20-1, 20-2, and 20-3, a PWM control unit 30 thatinputs voltages va, vb, and vc from the PI control unit 21, and aninverter 31 that converts a direct current into an alternating current.Current detecting circuits 32-1, 32-2, and 32-3, which detect the motorcurrents Ia, Ib, and Ic, are arranged between the inverter 31 and themotor 1. A feedback control system B, in which the detected motorcurrents Ia, Ib, and Ic are fed back to the subtracters 20-1, 20-2, and20-3, respectively, is formed.

Next, the current command value calculating unit 100 will be explained.First, concerning inputs thereof, a torque command value Tref calculatedfrom a torque detected by a not-shown torque sensor, a rotation angle θeof a rotor in the motor 1 detected by a position detecting sensor 11connected to the motor 1, and an electrical angular velocity ωecalculated by a differentiating circuit 24 are inputted. A convertingunit 101 calculates counter-electromotive forces ea, eb, and ec with theelectrical angular velocity ωe and the rotation angle θe of the rotor asinputs. Next, a three-phase/two-phase converting unit 102 converts thecounter-electromotive forces ea, eb, and ec into a d-axis componentvoltage ed and a q-axis component voltage eq. A q-axis command currentcalculating unit 108 calculates a current command value on a q-axisIqref with the d-axis component voltage ed and the q-axis componentvoltage eq as inputs. However, in this case, a current command value ona d-axis Idref is calculated as 0. In other words, in the followingoutput equation of a motor,Tref×ωm=3/2(ed×Id+eq×Iq)   (1)when Id=Idref=0 is inputted, the equation is calculated as follows.Iq=Iqref=2/3(Tref×ωm/eq)   (2)Phase current command values Iavref, Ibvref, and Icvref are calculatedon the basis of a current command value Iqref from the q-axis commandcurrent calculating unit 108 and an advance angle Φ of advance anglecontrol described later. In other words, a two-phase/three-phaseconverting unit 109 calculates the phase current command values Iavref,Ibvref, and Icvref based on the advance angle Φ calculated in theadvance angle calculating unit 107 and the current command value Iqrefcalculated in the q-axis command current calculating unit 108.

Note that a function such as Φ=a cos(ωb/ωm) or Φ=K(1−(ωb/ωm)) is usedempirically (“a cos” represents cos⁻¹). In addition, a motor baseangular velocity ωb is a motor limit angular velocity at the time whenthe motor 1 is driven without using field-weakening control. FIG. 2shows a relation between a torque T and a motor speed n (the angularvelocity ωe) and shows an example of the limit angular velocity ωb inthe case in which there is no field-weakening control.

Next, the advance angle control will be explained.

While the motor 1 is not rotating at high speed, that is, while amechanical angular velocity ωm of the motor 1 is lower than the motorbase angular velocity ωb, it is possible to output a torque complyingwith the torque command value Tref if the phase current command valuesIavref, Ibvref, and Icvref in accordance with a value calculated fromthe current command value Iqref by the two-phase/three-phase convertingunit 109 regardless of the advance angle Φ. This means that, as theelectric power steering apparatus, wheel operation by a driver isexecuted smoothly.

However, when the motor 1 rotates at high speed, that is, the mechanicalangular velocity ωm of the motor is higher than the motor base angularvelocity ωb, an angular velocity higher than the base angular velocityωb cannot be realized unless control taking into account the advanceangle Φ is executed. When this high-speed rotation of the motor 1 isconsidered from a viewpoint of the electric power steering apparatus, inthe case of sudden steering of a wheel for turn in parking a car oremergency shelter, steering feeling of the driver is deterioratedbecause the motor 1 does not follow the wheel operation.

There is a control system called field-weakening control as torquecontrol at the time of high-speed rotation of a motor. There is advanceangle control as a specific method of the field-weakening control.Details of this advance angle control system are described in U.S. Pat.No. 5,677,605 (C1) and C. C. Chan et al “Novel Permanent Magnet MotorDrivers for Electric Vehicles” IEEE Transaction on Industrialelectronics (Vol 43, No. 2 April 1996, page 335, FIG. 5). Acharacteristic of the advance angle control system is to advance a phaseof the current command value Iqref by the angle Φ to create afield-weakening component. In FIG. 10(B), when the current command valueIqref is advanced by the angle Φ, Iqref×sin Φ is generated as a d-axiscomponent and Iqref×cos Φ is generated as a q-axis component. Here,Iqref×sin Φ acts as a field-weakening component and Iqref×cos Φ acts asa torque component.

As a motor drive control system used in the electric power steeringapparatus, vector control, which is adapted to generate a rotatingmagnetic field from a control device via an inverter on the basis ofrotating position of a rotor to control to drive rotation of the rotor,is adopted. In other words, the vector control is adapted to, in pluralexciting coils arranged at intervals of a predetermined angle on anouter peripheral surface of the rotor, control rotation drive for therotor by sequentially switching excitation of the respective excitingcoils using a control circuit according to a rotor position.

This type of vector control is disclosed in, for example,JP-A-2001-18822. FIG. 3 is a block diagram showing an example of drivecontrol for a motor 56 according to the vector control.

In FIG. 3, a main path of a command signal leading to the motor 56 froma command current determining unit 51, which determines a controlcommand value of the motor 56, via a PI control unit 52, atwo-phase/three-phase coordinate converting unit 53, a PWM voltagegenerating unit 54, and an inverter 55 is formed. Current sensors 571and 572 are arranged between the inverter 55 and the motor 56. Afeedback path, in which a three-phase/two-phase coordinate convertingunit 59 converts a motor current detected by the current sensors 571 and572 into a two-phase current to feed back two-phase current componentsIq and Id to subtracting circuits 581 and 582 arranged between thecommand current determining unit 51 and the PI control unit 52, isformed.

With this control system, the command current determining unit 51receives the torque command value Tref calculated from a torque detectedby the torque sensor and a rotor rotating angle θ and an electricalangle ω detected by the position detecting sensor to determine currentcommand values Idref and Iqref. These current command values Idref andIqref are subjected to feedback correction by the two-phase currentcomponents Iq and Id, which are converted into two phases by thethree-phase/two-phase coordinate converting unit 59 in the feedbackpass, in the subtracting circuits 581 and 582, respectively. In otherwords, the subtracting circuits 581 and 582 calculate errors between thetwo-phase current components Id and Iq and the current command valuesIdref and Iqref. Thereafter, PI control units 521 and 522 calculatesignals, which indicate duty of PWM control, as command values Vd and Vqin forms of a d component and a q component. The two-phase/three-phasecoordinate converting unit 53 inversely converts the d component and theq component into three-phase components Va, Vb, and Vc. The inverter 55is subjected to the PWM control on the basis of the three-phase commandvalues Va, Vb, and Vc and an inverter current is supplied to the motor56 to control rotation of the motor 56.

Note that reference numeral 61 denotes a vehicle speed sensor; 62, asensitive area judging circuit; 63, a coefficient generating circuit;64, a basic assist force calculating circuit; 65, a return forcecalculating circuit; 66, an electrical angle converting unit; 67, anangular velocity converting unit; and 68, a non-interference controlcorrection value calculating unit.

In the case of the vector control described above, the current commandvalues Idref and Iqref are determined on the basis of the torque commandvalue Tref, the electrical angle •, and the rotation angle θ. Feedbackcurrents Iu and Iw of the motor 56 are converted into three-phasecurrents Iu, Iv, and Iw and, then, converted into two-phase currentcomponents Id and Iq. Thereafter, the subtracting circuits 582 and 581calculate errors between the two-phase current components Id and Iq andthe current command values Idref and Iqref. Current control by the PIcontrol is executed according to the errors, whereby command values Vdand Vq to the inverter 55 are calculated. Then, thetwo-phase/three-phase coordinate converting unit 53 inversely convertsthe command values Vd and Vq into the three-phase command values Va, Vb,and Vc, whereby the inverter 55 is controlled to perform drive controlfor the motor 56.

Incidentally, the d-axis component and the q-axis component generated bythe advance angle control simply advance the current command value Iqrefby the phase Φ. Thus, Iqref×sin Φ on the d-axis and Iqref×cos Φ on theq-axis are restricted to a fixed relation and a quantitative balance isnot always optimized. As a result, a motor terminal voltage is saturatedat the time of high-speed rotation and a motor current cannot follow acurrent command value, whereby torque ripple increases and motor noisealso increases. Therefore, as the electric power steering apparatus,inconveniences are caused in that, for example, a driver feels abnormalvibration through a wheel at the time of rapid wheel steering and motornoise is cased to give unpleasant feeling to the driver.

In the case of the vector control described above, a detection currentof the motor 56 and an output of the inverter 55 are in three phases andthe feedback control system is in two phases. It is necessary to controlto drive the motor 56 by inversely converting two phases into threephases in the two-phase/three-phase coordinate converting unit 53 inthis way. Thus, there is a problem in that the entire control system iscomplicated because the two-phase/three-phase conversion and thethree-phase/two-phase conversion are mixed.

In the control of the motor 56, if it is possible to maintain linearityof the control system, control responsiveness is improved. Thus, thecontrol is easy and a control target is easily attained. However,various nonlinear factors are included in drive control for the motor56. As a factor causing nonlinearity of motor drive, for example, thereis dead time of inverter control. Although an FET is used as a switchingelement of an inverter, the FET is not an ideal switching element. Inorder to prevent short circuit in upper and lower arms, a period forsetting both FETs of the upper and lower arms in an off state (deadtime) is provided. A nonlinear element of a switching transition stateis included in a motor current generated by switching of the FETs havingsuch dead time. In addition, a nonlinear element is also included in adetection element, a detecting circuit, and the like for detecting amotor current.

As a result, for example, a nonlinear element generated in an a-phasecurrent Ia is included in the d-axis current component Id and the q-axiscurrent component Iq by the d-q conversion in the three-phase/two-phasecoordinate converting unit 59 of the feedback system. Therefore, currentcontrol is performed on the basis of the current components Id and Iq,the command values Vd and Vq from the PI control units 522 and 521 tothe inverter 55 are calculated, a d-phase and a q-phase are inverselyconverted into an a-phase, a b-phase, and a c-phase in thetwo-phase/three-phase coordinate converting unit 53, and three-phasecommand values Va, Vb, and Vc are calculated. Consequently, thenonlinear element originally included in the a-phase current Ia isdiffused to the command values Va, Vb, and Vc of the inverter 55 by thed-q conversion. Thus, the nonlinear element is included in the commandvalues of not only the a-phase but also the b-phase and the c-phase. Inother words, in the case of the conventional control system, despite thefact that the motor is driven in three phases, feedback current controlis calculated in two phases and the command values Vd and Vq determinedin two phases are converted into the three-phase command values Va, Vb,and Vc formally and controlled. Thus, the nonlinear element is diffused.

Therefore, according to the conventional motor control, there is aproblem in that torque ripple is large and noise of the motor is alsolarge. When such motor control is applied to the electric power steeringapparatus, the electric power steering apparatus cannot assist steeringaccurately and smoothly following wheel operation. Thus, there is aproblem in that a driver feels vibration at the time of steering andnoise increases.

The invention has been devised because of the circumstances describedabove and it is an object of the invention to provide a motor and adrive control device for the motor, in which torque ripple is reducedand noise is reduced by controlling nonlinear elements included in motorcontrol in a state in which the nonlinear elements are separated intorespective phases, and also provides an electric power steeringapparatus that adopts the motor and the drive control device to have animproved steering performance and satisfactory steering feeling.

It is another object of the invention to provide a motor drive controldevice, in which a motor terminal voltage is not saturated even at thetime of high-speed rotation of a motor, torque ripple is reduced andmotor noise is reduced, and an electric power steering apparatus inwhich noise is reduced at the time of rapid steering of a wheel and withwhich wheel operation can follow the steering smoothly.

DISCLOSURE OF THE INVENTION

The invention relates to a motor and the above-described object of theinvention is attained by, when an induced voltage waveform is arectangular waveform of a motor or a pseudo-rectangular waveform and anorder wave component at the time when the rectangular waveform or thepseudo-rectangular waveform is subjected to frequency analysis isassumed to be n (=2, 3, 4, . . . ), setting the order wave component nequal to or larger than 5% of an amplitude component to satisfy thefollowing inequality:n×P/2×ω≦an upper limit value of a response frequency of current control

where P is the number of poles and ω is the number of actual rotation.

The invention relates to a motor drive control device that controls amotor having three or more phases. The motor drive control device has avector control phase current command value calculating unit thatcalculates phase current command values of the respective phases of themotor using vector control, a motor current detecting circuit thatdetects motor phase currents of the respective phases of the motor, anda current control unit that controls phase currents of the motor on thebasis of the phase current command values and the motor phase currents,whereby the object of the invention is attained. In addition, the vectorcontrol phase current command value calculating unit has acounter-electromotive force of each phase calculating unit thatcalculates a counter-electromotive force of each phase, a d-q voltagecalculating unit that calculates voltages ed and eq, which are d-axisand q-axis components of a counter-electromotive force, from thecounter-electromotive force of each phase, a q-axis command currentcalculating unit that calculates a current command value Iqref, which isa q-axis component of a current command value, from the voltages ed andeq, a d-axis command current calculating unit that calculates a currentcommand value Idref that is a d-axis component of a current commandvalue, and an each-phase current command calculating unit thatcalculates phase current command values of the respective phases fromthe current command values Iqref and Idref, whereby the object of theinvention is attained. Further, when the motor has three phases, phasecurrent command values Iavref, Ibvref, and Icvref are calculatedaccording to a constant depending on the current command values Idrefand Iqref and a rotation angle θe of the motor, whereby the object ofthe invention is attained.

The current control circuit includes integral control, the motor is abrushless DC motor, a current of the motor is a rectangular wave or apseudo-rectangular wave, or the motor drive control device is used in anelectric power steering apparatus, whereby the object of the inventionis attained more effectively.

Moreover, the invention relates to a motor drive control device thatcontrols a current of a motor on the basis of current command valuesIdref and Iqref, which are calculated using vector control. When adetected mechanical angular velocity ωm of the motor is higher than abase angular velocity ωb of the motor, the current command value Idrefis calculated according to a torque command value Tref of the motor, thebase angular velocity ωb, and the mechanical angular velocity ωm,whereby the object of the invention is attained.

The current command value Idref is calculated according to the torquecommand value Tref and a function of sin Φ and an advance angle Φ isderived from the base angular velocity ωb and the mechanical angularvelocity ωm, the current command value Iqref is calculated bysubstituting the current command value Idref in a motor output equation,or a motor current of the brushless DC motor is a rectangular wavecurrent or a pseudo-rectangular wave current, whereby the object of theinvention is attained more effectively.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a control block diagram based on conventional advance anglecontrol.

FIG. 2 is a graph showing a base angular velocity that is a limitangular velocity in the case in which field-weakening control is notused.

FIG. 3 is a block diagram showing a control system of conventionalvector control.

FIG. 4 is a sectional structure diagram showing an example of abrushless DC motor that is a control object of the invention.

FIG. 5 is a diagram showing a principle of rotor position detection.

FIG. 6 is a graph concerning an explanation of a definition of atrapezoidal wave current (voltage).

FIG. 7 is a graph showing an example of an induced voltage waveform (arectangular waveform).

FIG. 8 is a block diagram showing an example of a control system of abrushless DC motor according to the invention.

FIG. 9 is a block diagram showing an example of a structure forcalculation of a current command value Idref according tofield-weakening control of the invention.

FIG. 10 is a diagram showing a vector relation between current commandvalues Idref and Iqref according to a control system of the inventionand a conventional advance angle control system.

BEST MODE FOR CARRYING OUT THE INVENTION

An embodiment of the invention will be explained with reference to thedrawings.

In this embodiment, a three-phase brushless DC motor will be explained.However, the invention is not limited to this and can also be applied toother motors.

As shown in FIG. 4, a three-phase brushless DC motor 1 according to theinvention includes a cylindrical housing 2, a rotating shaft 4 that isdisposed along an axis of this housing 2 and supported to rotate freelyby bearings 3 a and 3 b at upper lower ends thereof, a permanent magnet5 for motor drive fixed to this rotating shaft 4, and a stator 6 that isfixed to an inner peripheral surface of the housing 2 to surround thispermanent magnet 5 and wound with three-phase excitation coils 6 a, 6 b,and 6 c. A rotor 7 is constituted by the rotating shaft 4 and thepermanent magnet 5. A ring-shaped permanent magnet 8 for phase detectionis fixed near an end of the rotating shaft 4 of this rotor 7. Thispermanent magnet 8 is polarized in an S pole and an N pole alternatelyat equal intervals in a peripheral direction.

A support base plate 10, which is made of a ring-shaped thin plate, isdisposed via a stay 9 on an end face on a side where the bearing 3 b isdisposed in the housing 2. A rotor position detector 11 such as aresolver or an encoder is fixed to this support base plate 10 so as tobe opposed to the permanent magnet 8. Note that, as shown in FIG. 5,actually, plural rotor position detectors 11 are provided to be spacedapart from each other appropriately in the peripheral direction inassociation with drive timing of the excitation coils 6 a to 6 c. Here,the excitation coils 6 a to 6 c are disposed to surround an outerperipheral surface of the rotor 7 while being separated from each otherby 120 degrees in an electrical angle. All coil resistances of therespective excitation coils 6 a to 6 c are set to be equal.

The rotor position detector 11 is adapted to output a position detectionsignal according to a magnetic pole of the opposed permanent magnet 8.The rotor position detector 11 is adapted to detect a rotating positionof the rotor 7 using a characteristic that the rotor position detector11 changes according to the magnetic pole of the permanent magnet 8. Inresponse to this rotating position, a vector control phase currentcommand value calculating unit 20 described later drives to rotate therotor 7 according to a two-phase excitation system for sequentiallyswitching the excitation coils 6 a to 6 c phase by phase while feedingan electric current to two phases of the three-phase excitation coils 6a to 6 c simultaneously.

Then, the drive control for the motor 1 is performed using a rectangularwave current or a pseudo-rectangular wave current as a motor current orusing a rectangular wave voltage or a pseudo-rectangular wave voltage asa motor induced voltage.

Here, the drive of the motor 1 is controlled according to therectangular wave current or the pseudo-rectangular wave current oraccording to the rectangular wave voltage or the pseudo-rectangular wavevoltage because, since the rectangular wave current or the rectangularwave voltage has a larger effective values compared with a sine wavecurrent or a sine wave voltage if a current peak value or a voltage peakvalue is the same, it is possible to obtain a larger output value(power). As a result, when motors having the same performance aremanufactured, there is an advantage that it is possible to realize areduction in size of a motor when the rectangular wave current or thepseudo-rectangular wave current is used as a motor current or therectangular wave voltage or the pseudo-rectangular wave voltage is usedas a motor induced voltage. On the other hand, there is a disadvantagethat it is difficult to reduce torque ripple according to control by therectangular wave current or the pseudo-rectangular wave current orcontrol by the rectangular wave voltage or the pseudo-rectangular wavevoltage of the induced voltage compared with control by the sine wavecurrent or the sine wave voltage.

FIG. 6 shows an example of a motor current waveform that is controlledaccording to current (Id) control. FIG. 6(A) shows a motor currentwaveform in the case in which the motor 1 rotates at relatively lowspeed and there is no field-weakening control according to the current(Id) control (Idref=0). FIG. 6(B) shows a motor current waveform in thecase in which the motor 1 rotates at high-speed and there isfield-weakening control according to the current (Id) control. FIG. 6(A)is a motor current waveform and a waveform of an induced voltagecorresponding to the motor current waveform is rectangular (trapezoidal)as shown in FIG. 7(A). An actual current waveform at the time of Id=0 isas shown in FIG. 7(B) (corresponding to FIG. 6(A)) as opposed to thewaveform of the induced voltage in FIG. 7(A). An actual current waveformat the time when Id=10[A] is as shown in FIG. 7(C) (corresponding toFIG. 6(B)) as opposed to the waveform of the induced voltage in FIG.7(A). Unlike a complete rectangular wave (a trapezoidal wave), therectangular wave current or the rectangular wave voltage referred to inthe invention includes a waveform having a recessed portion as shown inFIG. 6(A) or FIG. 7(B) or a waveform having a peak as shown in FIG. 6(B)or FIG. 7(C) or a current waveform (a pseudo-rectangular wave current)or a voltage waveform (a pseudo-rectangular wave voltage) shown in FIG.7(A).

The motor according to the invention is driven by an electric current ora voltage of n-th (=second, third, fourth, . . . ) harmonics. Afrequency of the n-th order is equal to or lower than an upper limitvalue (e.g., 1000 Hz) of a response frequency of current control. Inother words, when an induced voltage waveform of the motor is arectangular wave or a pseudo-rectangular wave and an order wavecomponent at the time when the rectangular waveform or thepseudo-rectangular waveform is subjected to frequency analysis isassumed to be n (=2, 3, 4, . . . ), the order wave component n equal toor larger than 5% of an amplitude component is represented by thefollowing expression (3):n×P/2×ω≦an upper limit value of a response frequency of current control  (3)

where P is the number of poles and ω is the number of actual rotation.

In this case, an angle sensor is provided such that a current waveformis provided at least as a function of the induced voltage waveform ofthe rectangular wave or the pseudo-rectangular wave. An electric timeconstant of motor correlation may be set to a control period or more oran angle estimating unit may be provided to give a motor currentwaveform at an estimated angle from this angle estimating unit.

A reason for setting the order wave component n equal to or larger than5% of an amplitude component according to expression (3) is as describedbelow. When the order wave component n, to which the current controlunit cannot respond, is on a current command value, the order wavecomponent n appears as torque ripple of the motor. It is known that, ifthe torque ripple of the motor is within 10%, the torque ripple iscontrolled by a torque control system so as not to be felt on a wheel(e.g., U.S. Pat. No. 3,298,006 (B2)). Therefore, it is possible todetermine a harmonic content of a counter-electromotive force such thatthe torque ripple is 10% or less in a current value (torque). A relationbetween the counter-electromotive force and the harmonic contentincluded in the current cannot be found uniquely according to a form ofvector control (or pseudo-vector control). However, it has been foundexperimentally that the torque ripple is 10% or less in a current value(torque) if the harmonic content is 5% or less of the amplitudecomponent.

In the electric power steering, usually, PWM control of 20 KHz isperformed. However, when a frequency is lower than 20 KHz, a problem ofmotor noise occurs and, when a frequency is higher than 20 KHz, aproblem of electromagnetic radiation noise and heat generation occurs.This depends on performance of an FET serving as driving means. In thePWM control of 20 KHz, 1000 Hz, which is 1/20 of 20 KHz, is an upperlimit value of a response frequency of the current control. In the PWMcontrol of 40 KHz, 2000 Hz, which is 1/20 of 40 KHz, is an upper limitvalue of a response frequency of the current control.

In the invention, a motor drive control device shown in FIG. 8 is formedfor the motor (with the number of poles P) having such characteristics.The motor drive control device of the invention includes a vectorcontrol phase current command value calculating unit 20, subtractingcircuits 20-1, 20-2, and 20-3 that calculates respective phase currenterrors on the basis of phase current command values Iavref, Ibvref, andIcvref from the vector control phase current instruction valuecalculating unit 20 and motor phase currents Ia, Ib, and Ic from currentdetecting circuits 32-1, 32-2, and 32-3, and a PI control unit 21 thatperforms proportional integral control. Respective phase commandcurrents are supplied from an inverter 31 to the motor 1 according toPWM control of the PWM control unit 30 to control rotation drive for themotor 1. An area A indicated by a broken line forms a current controlunit.

In this embodiment, in the vector control phase current command valuecalculating unit 20, current command values of vector control d and qcomponents are determined using an excellent characteristic of vectorcontrol and, then, the current command values are converted intorespective phase current command value. The vector control phase commandvalue calculating circuit 20 is closed with phase control rather than dand q control in a feedback control unit. Thus, since the theory ofvector control is used at a stage of calculating a current commandvalue, this control system is called pseudo-vector control (hereinafterreferred to as “PVC control”)

Note that the current control unit A in this embodiment includes thesubtracting circuits 20-1, 20-2, and 20-3 that calculate respectivephase current errors from the respective phase current command valuesIavref, Ibvref, and Icvref and the motor phase currents Ia, Ib, and Icof the motor 1 and the PI control unit 21 that inputs the respectivephase current errors. In addition, the current detecting circuits 32-1,32-2, and 32-3 are arranged as motor current detecting circuits betweenthe inverter 31 and the motor 1 to form a feedback current B thatsupplies the respective phase currents Ia, Ib, and Ic of the motordetected by the current detecting circuits 32-1, 32-2, and 32-3 to thesubtracting circuits 20-1, 20-2, and 20-3.

In addition, the vector control phase current command value calculatingunit 20 includes a converting unit 101 serving as acounter-electromotive force of each phase calculating unit, athree-phase/two-phase converting unit 102 serving as a d-axis and q-axisvoltage calculating unit, a q-axis command current calculating unit 103that calculates a current command value Iqref on a q-axis, atwo-phase/three-phase converting unit 104 serving as an each-currentcommand calculating unit, a d-axis command current calculating unit 105that calculates a current command value Idref on a d-axis, and aconverting unit 106 that converts a motor base angular velocity ωb fromthe torque command value Tref. The vector control phase current commandvalue calculating unit 20 receives a rotor position detection signal,which consists of a rotation angle θe of the rotor 7 detected by a rotorposition detector 11 such as a resolver and an electrical angularvelocity ωe obtained by calculating the rotation angle θe in thedifferential circuit 24, and a torque command value Tref determined onthe basis of a torque detected by a not-shown torque sensor andcalculates a phase command value signal according to the vector control.The rotor position detector 11 has a function as an angle sensor and itis possible to replace the rotor position detector 11 with an angleestimating unit.

The torque command value Tref is inputted to the q-axis command currentcalculating unit 103, the converting unit 106, and the d-axis commandcurrent calculating unit 105. The rotation angle θe is inputted to theconverting unit 101, the three-phase/two-phase converting unit 102, andthe two-phase/three-phase converting unit 104. The electrical angularvelocity ωe is inputted to the converting unit 101, the q-axis commandcurrent calculating unit 103, and the d-axis command current calculatingunit 105.

In such a monitor drive control device using the PVC control, drivecontrol for the motor 1 is performed as described below.

First, the vector control phase current command value calculating unit20 inputs the rotation angle θe and the electrical angular velocity ωeof the rotor 7 to the converting unit 101 and calculatescounter-electromotive forces ea, eb, and ec of respective phases on thebasis of a conversion table stored in the converting unit 101. Thecounter-electromotive forces ea, eb, and ec are a rectangular wave or apseudo-rectangular 605 wave of an n-th harmonics. A frequency of then-th harmonics is obtained by multiplying an electrical angular velocityof the motor by n. When an actual speed of the motor is assumed to be ω,an electrical angular velocity of the motor is represented by P/2×ω.Next, the three-phase/two-phase converting unit 102 610 serving as thed-q voltage calculating unit converts the counter-electromotive forcesea, eb, and ec into voltages ed and eq of d-axis and q-axis componentson the basis of the following expressions (4) and (5). $\begin{matrix}{\begin{bmatrix}{ed} \\{eq}\end{bmatrix} = {C\quad{1\begin{bmatrix}{ea} \\{eb} \\{ec}\end{bmatrix}}}} & (4) \\{{C\quad 1} = {\frac{2}{3}\begin{bmatrix}{- {\cos\left( {\theta\quad e} \right)}} & {- {\cos\left( {{\theta\quad e} - {2{\pi/3}}} \right)}} & {- {\cos\left( {{\theta\quad e} + {2{\pi/3}}} \right)}} \\{\sin\left( {\theta\quad e} \right)} & {\sin\left( {{\theta\quad e} - {2{\pi/3}}} \right)} & {\sin\left( {{\theta\quad e} + {2{\pi/3}}} \right)}\end{bmatrix}}} & (5)\end{matrix}$

Next, a method of calculating the current command value Idref on thed-axis, which is an important point of the invention, 620 will beexplained.

The d-axis command current calculating unit 105 calculates the d-axiscurrent command value Idref in accordance with the following expression(6) with the base angular velocity ωb from the converting unit 106, theelectrical angular velocity ωe from the differentiating circuit 24, andthe torque command value Tref from the torque sensor as inputs. Here, Ktis a torque coefficient and ωb is a motor base angular velocity. Theconverting unit 106 calculates the base angular velocity ωb with thetorque command value Tref as an input.Idref=−|Tref/Kt|·sin(a cos((ωb/ωm))   (6)

Concerning a term a cos(ωb/ωm) in expression (6), when rotating speed ofthe motor is not high, that is, when the mechanical angular velocity ωmof the motor 1 is lower than the base angular velocity ωb, since ωm<ωb,a cos(ωb/ωm)=0. Thus, Idref=0. When the rotating speed of the motor ishigh, that is, the mechanical angular velocity ωm is higher than thebase angular velocity ωb, a value of the current command value Idrefappears to start field-weakening control. As represented by expression(6), since the current command value Idref changes according to rotatingspeed of the motor 1, there is an excellent effect that it is possibleto perform control at the time of high-speed rotation seamlessly andsmoothly.

As another effect, there is also an effect concerning a problem ofsaturation of a motor terminal voltage. In general, a motor phasevoltage V is represented by the following expression.V=E+R·I+L(di/dt)   (7)Here, E is a counter-electromotive force, R is a fixed resistance, and Lis an inductance. The counter-electromotive force E increases as themotor rotates at higher speed. A power supply voltage such as a batteryvoltage is fixed. Thus, a range of a voltage, which can be used forcontrol of the motor, is narrowed. An angular velocity reaching voltagesaturation is the base angular velocity ωb. When the voltage saturationoccurs, a duty ratio of the PWM control reaches 100%. A motor currentcannot follow a current command value any more. As a result, torqueripple increases.

However, the current command value Idref represented by expression (6)has a negative polarity and an induced voltage component of the currentcommand value Idref concerning L(di/dt) of expression (7) has a polarityopposite to that of the counter-electromotive force E. Thus, there is aneffect that the counter-electromotive force E, which increases as themotor rotates at higher speed, is reduced by a voltage induced by thecurrent command value Idref. As a result, even if the motor 1 rotates athigh speed, the range of a voltage, which can control the motor, isincreased by the effect of the current command value Idref. In otherwords, there is an effect that a control voltage for the motor is notsaturated by the field-weakening control according to the control forthe current command value Idref, the range of a voltage, which cancontrol the motor, is increased, and torque ripple is prevented fromincreasing even at the time when the motor rotates at high speed.

FIG. 9 shows a block diagram of a circuit system concerning calculationof the current command value Idref. In FIG. 9, the torque command valueTref is inputted to the converting unit 106 and a torque coefficientunit 105 d. The motor electrical angular velocity ωe is inputted to themechanical angle calculating unit 105 a. The mechanical anglecalculating unit 105 a calculates a motor mechanical angular velocityωm(=ωe/P) from the motor electrical angular velocity ωe and inputs themechanical angular velocity in an a cos calculating unit 105 b. Theconverting unit 106 converts the torque command value Tref into the baseangular velocity ωb and inputs the base angular velocity ωb to the a coscalculating unit 105 b. The torque coefficient unit 105 d converts thetorque command value Tref into a coefficient Iqb(=Tref/Kt) and inputsthe coefficient to an absolute value unit 105 e. An aces calculatingunit 105 b calculates an advance angle Φ=a cos(ωb/ωm) on the basis ofthe inputted mechanical angular velocity ωm and base angular velocity ωband inputs the advance angle to a sin calculating unit 105 c. The sincalculating unit 105 c calculates sin Φ from the inputted advance angleΦ and inputs sin Φ to a multiplier 105 f for multiplying a value by −1.The multiplier 105 f multiplies the advance angle Φ from the sincalculating unit 105 c by an absolute value |Iqb| from an absolute valueunit 105 e and multiplies a result of the multiplication by −1 to obtainthe current command value Idref. The current command value Idef iscalculated according to expression (8) and is set as an output of thed-axis command current calculating unit 105.Idref=−|Iqb|×sin(a cos((ωb/ωm))   (8)

The current command value Idref calculated in accordance with expression(8) is inputted to the q-axis command current calculating unit 103 andthe two-phase/three-phase converting unit 104.

On the other hand, the q-axis command current calculating unit 103calculates the current command value Iqref on the q-axis on the basis ofthe two-phase voltages ed and eq, the electrical angular velocity ωe(=ωm×P), and the current command value Idref on the d-axis according toa motor output equation indicated by the following expressions (9) and(10).Tref×ωm=3/2(ed×Id+eq×Iq)   (9)Therefore, when Id=Idref and Iq=Iqref are substituted in expression (9),Iqref=2/3(Tref×ωm−ed×Idref)/eq   (10)In addition, a value calculated in expression (8) only has to besubstituted in the current command value Idref.

As indicated by expression (10), since the current command value Iqrefis derived from a motor output equation, which indicates that an outputof a motor is equivalent to electric power, it is possible to easilycalculate the current command value Iqref. In addition, it is possibleto calculate an optimum current command value Iqref balanced with thecurrent command value Idref for obtaining the necessary command torqueTref. Therefore, a motor terminal voltage is not saturated even at thetime when the motor is rotating at high speed and it is possible toperform control for minimizing torque ripple.

FIG. 10(A) illustrates a relation between the current command valuesIdref and Iqref in the invention described above. FIG. 10(B) shows arelation in the case of the conventional advance angle control system.

The current command values Idref and Iqref are inputted to thetwo-phase/three-phase converting unit 104 serving as the each-phasecurrent command value calculating unit and converted into the phasecurrent command values Iavref, Ibvref, and Icvref of the respectivephases. The current command values Idref and Iqref and the phase currentcommand values Iavref, Ibvref, and Icvref are represented likeexpressions (12) and (13). Here, a subscript, for example, “avref” ofthe phase current command value Iavref represents a phase currentcommand value of an a-phase determined by the vector control. Note that,as indicated by expression (13), a determinant C2 is a constant that isdetermined by the motor rotating angle θe. $\begin{matrix}{\begin{bmatrix}{Iavref} \\{Ibvref} \\{Icvref}\end{bmatrix} = {C\quad{2\begin{bmatrix}{Idref} \\{Iqref}\end{bmatrix}}}} & (12) \\{{C\quad 2} = \begin{bmatrix}{- {\cos\left( {\theta\quad e} \right)}} & {\sin\left( {\theta\quad e} \right)} \\{- {\cos\left( {{\theta e} - {2{\pi/3}}} \right)}} & {\sin\left( {{\theta\quad e} - {2{\pi/3}}} \right)} \\{- {\cos\left( {{\theta\quad e} + {2{\pi/3}}} \right)}} & {\sin\left( {{\theta\quad e} + {2{\pi/3}}} \right)}\end{bmatrix}} & (13)\end{matrix}$

Conventionally, the two-phase/three-phase converting unit 109 in FIG. 1calculates the phase current command values Iavref, Ibvref, and Icvrefusing the current command value Iqref and the advance angle Φ. In theinvention, as described above, the two-phase/three-phase converting unit104 calculates the phase current command values Iavref, Ibvref, andIcvref with the current command values Idref and Iqref as inputs. Then,the subtracting circuit 20-1, 20-2, and 20-3 subjects the respectivephase currents Ia, Ib, and Ic of the motor detected by the currentdetecting circuits 32-1, 32-2, and 32-3 and the phase current commandvalues Iavref, Ibvref, and Icvref to calculate respective errors. Next,the PI control unit 21 controls the errors of the respective phasecurrents to calculate a command value for the inverter 31, that is, thevoltage values va, vb, and vc representing duty of the PWM control unit30. The PWM control unit 30 subjects the inverter 31 to the PWM controlon the basis of the voltage values va, vb, and vc, whereby the motor 1is driven and a desired torque is generated.

As explained above, in the motor and the drive control device for themotor of the invention, a motor terminal voltage is not saturated evenat the time when the motor is rotating at high speed. Thus, control forminimizing torque ripple is possible. Therefore, when the invention isapplied to an electric power steering apparatus, it is possible toexecute rapid wheel steering smoothly. Thus, there is an excellenteffect that a driver is prevented from feeling a sense of incongruitydue to vibration of a wheel or the like.

The invention is completely different from the feedback controlaccording to the d and q control of the conventional technique in thatfeedback control is executed according to only control of the respectivephases. As a result, whereas, in the conventional technique, there is aproblem in that a nonlinear element caused in the a-phase disperses tothe respective phases b and c in a process of executing the conventionalfeedback control according to the d and q control and correction controlcannot be performed correctly, in the invention, since a nonlinearelement of the a-phase is subjected to the feedback control only in thea-phase and is not distributed to the b-phase and the c-phase,correction control can be performed correctly.

By using such PVC control, it is possible to control a motor in a statein which a nonlinear element included in control is separated intorespective phases. As a result, it is possible to realize motor controlwith reduced torque ripple and reduced noise. Therefore, when theinvention is applied to an electric power steering apparatus, it ispossible to perform smooth wheel operation with reduced noise andreduced vibration at the time of parking and in emergency steering.

Note that, although the phase voltages ea, eb, and ec are used in theembodiment, the same effect is obtained when the phase voltages areconverted into line voltages eab, ebc, and eca, or the like to controlthe motor.

As described above, according to the motor of invention, there is aneffect that a motor terminal voltage is not saturated even at the timewhen the motor is rotating at high speed, torque rippled is reduced, andmotor noise is reduced. Moreover, in the electric power steeringapparatus of the invention, there is an excellent effect that it ispossible to provide an electric power steering apparatus that followsrapid steering of a wheel smoothly, does not cause a sense ofincongruity in wheel operation, and has reduced noise.

In addition, according to the electric power steering apparatusaccording to the invention, respective phase current command values arecalculated on the basis of the vector control, and the PVC control forcontrolling respective phases separately is used as current feedbackcontrol. Thus, it is possible to provide a motor drive control devicethat can control the brushless DC motor to be small, have reduced torqueripple, and have reduced motor noise. It is possible to provide anelectric power steering system with which wheel operation is smooth andnoise is reduced.

Moreover, according to the motor of the invention, a frequency of n-thharmonics is equal to or lower than an upper limit value of a responsefrequency of current control. Thus, even if the motor is driven with arectangular wave current or a pseudo-rectangular wave current or drivenwith a rectangular wave voltage or a pseudo-rectangular wave voltage,the motor has reduced torque ripple, a reduced size, and reduced noise.

INDUSTRIAL APPLICABILITY

According to the invention, a motor terminal voltage is not saturatedeven at the time when a motor is rotating at high speed, toque ripple isreduced, and motor noise is reduced. Thus, if the invention is appliedto an electric power steering apparatus, it is possible to provide anelectric power steering apparatus that follows rapid steering of a wheelsmoothly, does not cause a sense of incongruity in wheel operation, andhas reduced noise.

According to the electric power steering apparatus in the invention,respective phase current instruction values are calculated on the basisof the vector control, and the PVC control for controlling respectivephases separately is used as current feedback control. Thus, it ispossible to provide a motor drive control device that can control thebrushless DC motor to be small, have reduced torque ripple, and havereduced motor noise. It is possible to provide an electric powersteering system with which wheel operation is smooth and noise isreduced.

1. A motor drive control device that controls a motor having threephases, wherein a waveform of a current or an induced voltage of themotor is a rectangular wave or a pseudo-rectangular wave having n-th(n=2, 3, 4, . . . ) harmonics, comprising: a d-q voltage calculatingunit that calculates a voltage ed and a voltage eq, which arerespectively d-axis and q-axis components of a counter-electromotiveforce; a d-axis command current calculating unit that calculates acurrent command value Idref that is a d-axis component of the currentcommand value on the basis of a torque command value Tref of the motor,a base angular velocity ωb of the motor, and a detected mechanicalangular velocity ωm of the motor, when the mechanical angular velocityωm is higher than the base angular velocity ωb; a q-axis command currentcalculating unit that calculates a current command value Iqref, which isa q-axis component of a current command value, on the basis of thevoltage ed, the voltage eq, and the current command value Idref; and acurrent control unit that controls currents of the motor on the basis ofthe current command value Idref and the current command value Iqref. 2.A motor drive control device according to claim 1, wherein the currentcommand value Iqref is calculated by substituting the mechanical angularvelocity ωm, the voltage ed, the voltage eq, the torque command valueTref, and the current command value Idref into a motor output equation.3. A motor drive control device according to claim 1 or 2, wherein themotor is a brushless DC motor.
 4. An electric power steering apparatus,wherein the motor drive control device according to claim 1 or 2 isprovided.
 5. An electric power steering apparatus, wherein the motordrive control device according to claim 3 is provided.